Converter digital control circuit with adaptive feedforward compensation

ABSTRACT

A control circuit for a voltage regulator includes a divider coupled to the regulated output voltage to generate a divided voltage having a value that is a fraction of the regulated output voltage, an ADC responsive to the divided voltage to generate a feedback control signal, a digital compensator responsive to the feedback control signal and to a feedforward control signal scaled by a feedforward gain value to generate a compensator signal, and a pulse width modulator responsive to the compensator signal to generate a voltage control signal to control a switch of the voltage regulator. The digital compensator includes a register configured to store a value indicative of the input voltage and a feedforward gain unit configured to generate the feedforward gain value in response to the value indicative of the input voltage. In embodiments, the feedforward gain value is generated in response to the square of the value indicative of the input voltage. The feedforward gain value can be updated each time the value indicative of the input voltage is updated.

CROSS REFERENCE TO RELATED APPLICATIONS

Not Applicable.

STATEMENT REGARDING FEDERALLY SPONSERED RESEARCH

Not Applicable.

FIELD

This disclosure relates generally to control circuits, and more particularly, to adaptive feedforward compensation for digitally controlled voltage mode DC-DC converters.

BACKGROUND

As is known, DC-DC converters, such as buck converters, boost converters, and other converter types, often use control circuitry and techniques to control a level of one or more signals of the converters (e.g., voltage output signals). The control circuitry can be implemented with digital and/or analog circuitry and techniques. In digital control, control signals generated by feedback and/or feedforward control circuitry of a converter may be used to generate a duty cycle word indicative of a required duty cycle for generating a desired converter output voltage. The duty cycle word may be converted into a voltage control signal for controlling a power stage of the converter from which the converter output voltage is generated.

Compensation circuitry and techniques for the digital control loop are generally tailored to a particular set of operating conditions. For example, control parameters such as loop bandwidth and crossover frequency may be selected based on a desired operating frequency and converter output voltage. If operating conditions change, it may be necessary to set new parameter values.

SUMMARY

Described herein are concepts, systems, circuits and techniques related to a control circuit for a voltage regulator including a digital compensator configured to perform adaptive feedforward compensation to permit operation over different operating conditions in a manner that responds more effectively to input voltage transients. In one aspect of the concepts described herein, the digital compensator is responsive to a feedback control signal and to a feedforward control signal scaled by a feedforward gain value to generate a compensator signal. The digital compensator can include a register configured to store a value indicative of the input voltage and a feedforward gain unit configured to generate the feedforward gain value in response to the value indicative of the input voltage. In embodiments, the feedforward gain value is generated in response to the square of the value indicative of the input voltage. The feedforward gain value can be updated each time the value indicative of the input voltage is updated.

The regulator includes at least one switch responsive to a voltage control signal for switching at a switching frequency and an output filter having a corner frequency, the voltage regulator configured to convert an input voltage into a regulated output voltage. The regulator further includes a divider coupled to the regulated output voltage to generate a divided voltage having a value that is a fraction of the regulated output voltage, an ADC responsive to the divided voltage to generate a feedback control signal, and the digital compensator. The compensator is responsive to the feedback control signal to generate a compensator signal and a pulse width modulator is responsive to the compensator signal to generate the voltage control signal. An error amplifier may be responsive to the divided voltage and to a reference voltage to generate an error voltage for coupling to the ADC.

The control circuit may include one or more of the following features individually or in combination with other features. The digital compensator may include a proportional-integral-derivative (PID) controller and the register configured to store a value indicative of the input voltage can be an integral register. One or more of a proportional gain, an integral gain, and a derivative gain of the PID controller is scaled by the value indicative of the input voltage. The feedforward control signal can include one or more of positive feedforward control signals and negative feedforward control signals. The feedforward gain value can be updated in response to the value indicative of the input voltage being updated and, in some embodiments, can be updated each time the value indicative of the input voltage is updated. The value indicative of the input voltage can be updated (1) each time one of the positive feedforward control signal or negative feedforward control signal is received; and (2) at an end of a clock cycle corresponding to the switch frequency. The digital compensator can include an accumulator configured to accumulate the positive feedforward control signals and negative feedforward control signals occurring during a clock cycle corresponding to the switching frequency, wherein the compensator signal is updated at the end of the clock cycle based on the positive feedforward control signals and negative feedforward control signals accumulated during the clock cycle. The digital compensator can be configured to generate the compensator signal during a first portion of a clock cycle corresponding to the switching frequency and the gain of one or more of a proportional gain, an integral gain, and a derivative gain of the PID controller can be scaled by the value indicative of the input voltage during a second portion of the clock cycle different than the first portion of the clock cycle. The voltage regulator can be a DC-DC converter.

Also described is a method for generating a voltage control signal for controlling a switch of a voltage regulator configured to convert an input voltage into a regulated output voltage, the switch operating at a switching frequency, including generating a feedback control signal based on a voltage difference between the regulated output voltage and a reference voltage, generating a compensator signal with a digital compensator in response to the feedback control signal and in response to a feedforward control signal scaled by a feedforward gain value, wherein the feedforward control signal comprises one or more of positive feedforward control signals and negative feedforward control signals, and converting the compensator signal into the voltage control signal with a pulse width modulator.

The method for generating a control voltage may include one or more of the following features individually or in combination with other features. Generating the compensator signal comprises providing a digital compensator with a Proportional-Integral-Derivative (PID) controller and the method may further include scaling one or more of a proportional gain, an integral gain, or a derivative gain of the PID controller by the value indicative of the input voltage. Providing the digital compensator with a PID controller may include providing an integral register in which a value indicative of the input voltage is stored and the method may further include computing the feedforward gain value in response to the value indicative of the input voltage. Computing the feedforward gain value may include computing the feedforward gain value in response to the square of the value indicative of the input voltage. Computing the feedforward gain value may include updating the feedforward gain value each time the value indicative of the input voltage is updated. The value indicative of the input voltage may be updated (1) each time one of the positive feedforward control signal or negative feedforward control signal is received; and (2) at an end of a clock cycle corresponding to the switch frequency. The method may further include accumulating the positive feedforward control signals and negative feedforward control signals occurring during a clock cycle corresponding to the switching frequency, wherein the compensator signal is updated at the end of the clock cycle based on the positive feedforward control signals and negative feedforward control signals accumulated during the clock cycle.

In a further aspect of the concepts described herein, a control circuit for a voltage regulator including at least one switch responsive to a voltage control signal for switching at a switching frequency and configured to convert an input voltage into a regulated output voltage includes a divider coupled to the regulated output voltage to generate a divided voltage having a value that is a fraction of the regulated output voltage, an ADC responsive to the divided voltage to generate a feedback control signal, means, responsive to the feedback control signal and to a feedforward control signal scaled by a feedforward gain value, for generating a compensator signal, and a pulse width modulator responsive to the compensator signal to generate the voltage control signal.

The control circuit may include one or more of the following features individually or in combination with other features. The compensator signal generating means may include means, responsive to a value indicative of the input voltage, for generating the feedforward gain value. The feedforward gain value generating means may include means for updating the feedforward gain value when the value indicative of the input voltage is updated.

The control circuit includes a divider coupled to the regulated output voltage to generate a divided voltage having a value that is a fraction of the regulated output voltage, an error amplifier responsive to the divided voltage and to a reference voltage to generate an error voltage indicative of a difference between the divided voltage and the reference voltage, an ADC responsive to the error voltage to generate a feedback control signal, a digital compensator responsive to the feedback control signal to generate a compensator signal, and a pulse width modulator responsive to the compensator signal to generate the voltage control signal. The corner frequency of the output filter has a first fixed, predetermined relationship with respect to the switching frequency, a second fixed, predetermined relationship with respect to the crossover frequency, and a third fixed, predetermined relationship with respect to the at least one pole and at least one zero.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing features may be more fully understood from the following description of the drawings. The drawings aid in explaining and understanding the disclosed technology. Since it is often impractical or impossible to illustrate and describe every possible embodiment, the provided figures depict one or more illustrative embodiments. Accordingly, the figures are not intended to limit the scope of the broad concepts, systems and techniques described herein. Like numbers in the figures denote like elements.

FIG. 1 is a block diagram of an example regulator circuit that includes a control circuit for generating a feedforward control signal and including a digital compensator with adaptive compensation;

FIG. 2 is a block diagram of an example feedforward control circuit of the regulator circuit of FIG. 1;

FIG. 3 shows illustrative signal waveforms of various example input and output signals of the control circuit of FIG. 2;

FIG. 4 shows a small signal control loop representation of a voltage mode buck converter including a digital compensator with adaptive compensation;

FIG. 5 is a block diagram of an example PID controller of the digital compensator of FIG. 1;

FIG. 6 shows graphs illustrating the output signal response to input voltage transients and also shows the feedforward control signals;

FIG. 7 is a block diagram of a further example PID controller of the digital compensator; and

FIG. 8 is a block diagram of another example feedforward control circuit.

DETAILED DESCRIPTION

The features and other details of the concepts, systems, and techniques sought to be protected herein will now be more particularly described. It will be understood that any specific embodiments described herein are shown by way of illustration and not as limitations of the disclosure and the concepts described herein. Features of the subject matter described herein can be employed in various embodiments without departing from the scope of the concepts sought to be protected. Embodiments of the present disclosure and associated advantages may be best understood by referring to the drawings, where like numerals are used for like and corresponding parts throughout the various views.

Referring now to FIG. 1, a simplified view of an example buck regulator circuit 100 which may be provided in a power management integrated circuit (IC), for example, is shown. The regulator circuit 100 (e.g., a voltage mode regulator) includes a power stage 110 and a control circuit 120 and has an input 100 a at which an input or supply voltage V_(IN) is received and an output 100 b at which a regulated output voltage V_(OUT) (e.g., a “stepped-down” output voltage) is generated.

The power stage 110, which is illustrative of one example configuration of a regulator power stage, includes a buffer 111, an inverter 112, a first transistor 113 (e.g., a first power switch) and a second transistor 114 (e.g., a second power switch). The power stage 110 also includes an output resistor R_(out) and output filter (e.g., an LC filter) including an inductor L_(x) and a capacitor C_(BP). The inductor L_(x) has a DC resistance R_(L) and the capacitor C_(BP) has an equivalent series resistance R_(c), which are shown in the figure. In embodiments in which the buck regulator circuit 100 is provided in the form of an IC, it will be appreciated that various components may be integrated into the IC or may be external to the IC.

Transistors 113, 114 (e.g., field effect transistors (FETs)) are provided in a push-pull configuration in the illustrated embodiment, each having a corresponding first terminal 113 a, 114 a (e.g., a gate terminal) coupled to a node 110 a (here, a control node) of power stage 110 at which a voltage control signal for controlling a voltage level of the output voltage V_(OUT) is provided. The first terminals 113 a, 114 a of transistors 113, 114 are coupled to the node 110 a via buffer 111 and inverter 112, respectively. Transistor 113 has a second terminal 113 b (e.g., a drain terminal) coupled to regulator input 100 a at which the input voltage or supply voltage V_(IN) is provided and a third terminal 113 c (e.g., a source terminal) coupled to a node 110 b of power stage 110. Additionally, transistor 114 also has a second terminal 114 b (e.g., a drain terminal) coupled to node 110 b and a third terminal 114 c (e.g., a source terminal) coupled to a node 110 c of power stage 110, which node is coupled to a reference potential (here, GND).

Inductor L_(x) has a first terminal 115 a coupled to the node 110 b and a second terminal 115 b coupled to output voltage V_(OUT) node 100 b of power stage 110. Additionally, capacitor C_(BP) has a first terminal 116 a coupled to output node 100 b and a second terminal coupled to node 110 c of power stage 110, which node is coupled to GND. Further, output resistor R_(out) has a first terminal 117 a coupled to node 100 b and a second terminal 117 b coupled to GND node 110 c.

The control circuit 120 (e.g., a voltage mode control circuit), which generates the voltage control signal received at node 110 a of the power stage 110, as will be discussed further below, includes a feedback path 130 and a feedforward path 170. The control circuit 120 also includes a compensator 180 coupled to the feedback path 130 and to the feedforward path 170 and a digital Pulse Width Modulator (PWM) 190 coupled to the digital compensator 180.

The feedback path 130 includes a divider circuit 140, an error amplifier 150 and an analog-to-digital converter (ADC) 160. The divider 140 is coupled to regulator output 100 b and includes a first resistor R₁ and a second resistor R₂. Resistor R₁ has a first terminal 141 a coupled to the regulator output 100 b and a second terminal 141 b coupled to a node 140 a (i.e., an intermediate node) of the divider circuit 140 at which a divided voltage V_(DIV) is provided. The divided voltage V_(DIV) has a value that is a fraction of the output voltage V_(OUT) at the regulator output 100 b. Resistor R₂ of the divider 140 has a first terminal 142 a coupled to node 140 a and a second terminal 142 b coupled to a reference potential (here, GND).

The error amplifier 150 of the feedback path 130 is coupled to receive the divided voltage V_(DIV) at a first input 150 a (e.g., an inverting input) and a reference voltage V_(REF) at a second input 150 b (e.g., a non-inverting input) and is configured to generate an error voltage at an output 150 c in response to a difference between the first amplifier input 150 a and the second amplifier input 150 b. Additionally, the ADC 160 of the feedback path 130 is coupled to receive the error voltage at an ADC input 160 a and is configured to generate a converted digital signal (e.g., a binary word) e[n] at an ADC output 160 b in response to the ADC input 160 a. The converted digital signal e[n] corresponds to a feedback control signal of the feedback path 130 in the illustrated embodiment. It will be appreciated that in some embodiments, the error amplifier 150 and/or the resistor divider 140 may be eliminated in which case the output voltage or divided version thereof may be directly coupled to the ADC 160.

The feedforward path 170 of the control circuit 120, which may be referred to herein alternatively as the feedforward control circuit or simply control circuit 170, has a feedforward path input 170 a coupled to the input voltage V_(IN) received at input 100 a of the regulator circuit 100. The feedforward path 170 is configured to generate a first feedforward signal PT indicative of a positive transient of the input voltage V_(IN) at a first feedforward path output 170 b in response to the feedforward path 170 detecting that the input voltage V_(IN) experiences a positive transient (i.e., a voltage increase). Additionally, the feedforward path 170 is configured to generate a second feedforward signal NT indicative of a negative transient of the input voltage V_(IN) at a second feedforward path output 170 c in response to the feedforward path 170 detecting that the input voltage V_(IN) experiences a negative transient (i.e., a voltage decrease).

The compensator 180 (e.g., a digital compensator) of the control circuit 120 is coupled to receive the feedback control signal e[n] at a first compensator input 180 a. Additionally, the compensator 180 is coupled to receive the first feedforward signal PT at a second compensator input 180 b when the feedforward path 170 detects that the input voltage V_(IN) experiences a positive transient and to receive the second feedforward signal NT at a third compensator input 180 c when the feedforward path 170 detects that the input voltage V_(IN) experiences a negative transient. The compensator 180 is configured to generate a duty cycle word d[n] at a compensator output 180 d in response to the compensator inputs 180 a, 180 b, and 180 c. In general, the first feedforward signal PT will tend to cause the compensator output duty cycle word d[n] to reduce the duty cycle of the switches 113, 114 (in response to a positive input voltage variation or transient) and the second feedforward signal NT will tend to cause the compensator output duty cycle word d[n] to increase the duty cycle of the switches 113, 114 (in response to a negative input voltage variation or transient).

More particularly, in the illustrated embodiment the compensator 180 includes a proportional-integral-derivative (PID) controller 182, a feedforward duty cycle circuit 184 and a compensator output module 186. The PID controller 182 is coupled to receive the feedback control signal e[n] from the first compensator input 180 a at a PID controller input 182 a and is configured to generate a first duty cycle word d_(PlD)[n] associated with the feedback control signal e[n] at a PID controller output 182 b. Additionally, the feedforward duty cycle circuit 184 is coupled to receive the first feedforward signal PT and the second feedforward signal NT from the second and third compensator inputs 180 b, 180 c at first and second feedforward circuit inputs 184 a, 184 b, respectively, and is configured to generate a second duty cycle word d_(FFwd)[n] associated with the received first or second feedforward signals at a feedforward circuit output 184 c. Further, the compensator output module 186, which includes a summing circuit in the illustrated embodiment and may include a subtractor circuit in other embodiments, for example, is coupled to receive the first duty cycle word d_(PID)[n] at a first compensator output module input 186 a and the second duty cycle word d_(FFwd)[n] at a second compensator output module input 186 b and is configured to generate a third duty cycle word d[n] as a combination of the first and second duty cycle words d_(PID)[n], d_(FFwd)[n]. The third duty cycle word d[n] is provided to the compensator output 180 d.

In some embodiments, the PID controller 182 may contain fixed, constant gain parameters for the proportional, integral, and derivative components. In other embodiments however, according to a further aspect described below in connection with FIGS. 4 and 5, the PID controller 182 may be configured to perform adaptive compensation by mathematically computing adaptive gain parameters for use in order to thereby permit stable operation for different operating conditions. And according to another aspect, as described below in connection with FIG. 7 for example, additional adaptive compensation may be applied to a feedforward gain feedforward gain value.

The digital PWM circuit 190 of the control circuit 120 is coupled to receive the duty cycle word d[n] (i.e., the third duty cycle word d[n]) generated by the compensator 180 at a PWM input 190 a and is configured to generate a PWM, or voltage control signal at a PWM output 190 b. The voltage control signal has a duty cycle based on the duty cycle word d[n].

Transistors 113, 114 of power stage 110 are each coupled to receive the voltage control signal or an inverted version of the voltage control signal at a corresponding input 113 a, 114 a and are switched on and off (i.e., between conducting and non-conducting states) in response to transitions of the voltage control signal. More particularly, power stage 110 is coupled to receive the voltage control signal at node 110 a and is configured to generate two complementary transistor drive signals for controlling transistors 113, 114. It will be understood that the first and second transistor drive signals may be level shifted with respect to the voltage control signal at node 110 a and may be processed to introduce a dead time during which neither transistor 113, 114 is on.

Referring now to FIG. 2, an example feedforward control circuit 200 for generating a feedforward control signal according to an embodiment of the disclosure is shown. The control circuit 200, which may provide the feedforward path 170 in the regulator circuit 100 of FIG. 1, includes a first divider 210, a first buffer 220, a first capacitor 230, a current mirror circuit 240 and a digitizing circuit 260 from which the feedforward control signal is generated. More particularly, the control circuit 200 receives an input voltage V_(IN) that may be the same or similar to input voltage V_(IN) of FIG. 1 at an input 200 a of the control circuit 200 and generates a feedforward control signal based on the input voltage V_(IN). In the example circuit 200, the feedforward control signal may take the form of a first control signal NT provided at a first feedforward output 200 c and/or a second control signal PT provided at a second feedforward output 200 d of the control circuit 200.

The first divider 210 of the control circuit 200 includes a first resistor R₁ and a second resistor R₂. Resistor R₁ has a first terminal 211 a coupled to the input voltage V_(IN) and a second terminal 211 b coupled to a node 210 a (i.e., an intermediate node) of the divider 210 at which a divided voltage V_(DIV) having a value that is a fraction of the input voltage V_(IN) is generated. Additionally, resistor R₂ has a first terminal 212 a coupled to node 210 a and a second terminal 212 b coupled to a reference potential (here, ground or GND). Resistance values of the resistors Ri, R₂ may be selected to achieve a desired divided voltage VDIV, which divided voltage V_(DIV) is coupled to the first buffer 220. In particular, resistance values of the resistors R₁, R₂ may be selected to divide the input voltage V_(IN) by an amount sufficient to allow lower voltage rated components to be used in the control circuit 200, particularly in applications in which the input voltage V_(IN) can vary significantly, such as in automotive applications in which the input voltage can vary between about 10V and 60V. Resistors R₁ and R₂ may be selected to have relatively large resistance values in order to reduce static current drawn from the input voltage V_(IN).

The first buffer 220 may be a unity gain buffer configured to provide a buffered voltage at a buffer output 220 c in response to the divided voltage V_(DIV). The first capacitor 230 has a first terminal coupled to the buffer output 220 c and a second terminal 230 b, and a feedforward current I_(FFWD) flows through capacitor 230 when there is a variation (i.e., a positive or negative transient) in the input voltage V_(IN). More particularly, as will be described, under relatively steady state input voltage conditions there will be no current flow through capacitor 230 because there is no voltage drop across its terminals 230 a, 230 b. However, when the input voltage V_(IN) varies by more than a predetermined amount (i.e., experiences a predetermined positive or negative variation, referred to herein alternatively as the occurrence of a positive or negative input voltage transient), then current I_(FFWD) is generated. More particularly, if the input voltage transient is positive (i.e., the input voltage rises), then the feedforward current I_(FFWD) flows from the buffer output 220 c to the current mirror circuit 240; whereas if the input voltage transient is negative (i.e., the input voltage falls), then the feedforward current I_(FFWD) flows from the current mirror circuit 240 to the buffer output 220 c. Buffer 220 separates resistors R₁ and R₂ from capacitor 230, thereby advantageously reducing the time constant associated with the feedforward current IFFWD.

The current mirror circuit 240 has a first current path 241 coupled to the second capacitor terminal 230 b and to a first reference current source 243 and a second current path 244 coupled to a second reference current source 246. Additionally, the current mirror circuit 240 has a current mirror output node 240 c at which a current mirror output voltage (V_(RAMP)) indicative of the input voltage variation is generated. In embodiments, the first reference current source 243 and the second reference current source 246 are a same reference current source (i.e., the first current path and the second current path are coupled to a same current source), or at least provide substantially the same current level.

In the illustrated embodiment, the first current path 241 includes a first transistor 242 and the second current path 244 includes a second transistor 245. First transistor 242 (e.g., a FET) of the first current path 241 has a first terminal 242 a (e.g., a source terminal) coupled a node 240 a of the current mirror circuit 240, which node 240 a is coupled to the second terminal 230 b of the first capacitor 230 and to the first reference current source 243. First transistor 242 also has a second, control terminal 242 b (e.g., a gate terminal) coupled to a node 240 b of the current mirror circuit 240, which node 240 b is coupled to node 240 a of the current mirror circuit 240. Additionally, first transistor 242 has a third terminal 242 c (e.g., a drain terminal) coupled a reference potential (here, GND).

Second transistor 245 (e.g., a FET) of the second current path 244 has a first terminal 245 a (e.g., a source terminal) coupled to a node 240 c of the current mirror circuit 240 at which a current mirror output voltage indicative of a variation of the input voltage V_(IN) is generated, and coupled to the second reference current source 246. Additionally, second transistor 245 has a second, control terminal 245 b (e.g., a gate terminal) coupled to node 240 b and a third terminal 245 c (e.g., a drain terminal) coupled to the reference potential.

A second capacitor 250 has a first terminal 250 a coupled to a node 200 b of the control circuit 200, which node 200 b is coupled to node 240 c of the second current path at which the current mirror output voltage is generated. The second capacitor 250 also has a second terminal 250 b coupled to a reference potential, here GND. The current mirror output voltage is generated across the second capacitor 250.

In operation, if no feedforward current I_(FFWD) flows (i.e., as will occur when the input voltage level V_(IN) is relatively constant), then the same current provided by both of the current sources 243, 246 flows through the first and second current mirror circuit paths 241, 244. As a result, the voltage at nodes 240 a and 240 c will remain unchanged. If however, a positive input voltage transient occurs, causing a feedforward current I_(FFWD) to flow into current mirror node 240 a, then the voltage at node 240 a will rise and the voltage at current mirror output node 240 c will fall accordingly. The falling voltage at current mirror output node 240 c discharges the capacitor 250 causing the voltage across capacitor 250 to fall and this decrease in the current mirror output voltage V_(RAMP) will be sensed by the digitizing circuit 260 to provide an indication of the positive input voltage transient via the feedforward control signal PT. Conversely, if a negative input voltage transient occurs causing a feedforward current I_(FFWD) to flow out of current mirror node 240 a to the buffer output 220 c, then the voltage at node 240 a will fall and the voltage at current mirror output node 240 c will rise accordingly. The rising voltage at current mirror output node 240 c causes the capacitor 250 to charge and this increase in the current mirror output voltage V_(RAMP) will be sensed by the digitizing circuit 260 to provide an indication of the negative input voltage transient via the feedforward control signal NT. With the above explanation, it will be apparent that the same current that flows through capacitor 230 flows through capacitor 250, but with opposition polarity.

In embodiments, the first capacitor 230 has a first capacitance value and the second capacitor 250 has a second capacitance value that is substantially different from the first capacitance value. In embodiments, the first and second capacitance values are selected such that the current mirror output voltage generated at nodes 240 c, 200 b changes at a substantially same rate as the input voltage V_(IN). More particularly, since an equal but opposite polarity current flows through capacitor 250 as flows through capacitor 230, and since both capacitors 230, 250 are affected by the same rate of change of the input voltage dV_(IN)/dt, the rate of change of the voltage across both capacitors will be the same (albeit the rate of change of the voltage across capacitor 230 will be +dV_(IN)N/dt and the rate of change of the voltage across capacitor 250 will be −dV_(IN)/dt). As one of various examples, capacitor 230 may have a capacitance value of 50 pF and capacitor 250 may have a capacitance value of 5 pF.

The digitizing circuit 260, which is illustrative of one example configuration of a digitizing circuit according to the disclosure, includes a second divider 270, a first comparator 280 and a second comparator 290.

The second divider 270 of the digitizing circuit 260 includes a resistor ladder having a plurality of series-coupled resistors (here, resistors R₃, R₄, R₅, R₆). The divider 270 is coupled between a bandgap reference voltage V_(BG) and a reference potential, such as GND and has intermediate nodes 270 a, 270 b at which reference, or threshold voltages V+, V− are provided, respectively.

In some embodiments, the first threshold voltage V+ has a first voltage value and the second threshold voltage V− has a second voltage value that is different from the first voltage value. In other embodiments, the first voltage value of the first threshold voltage V+ is substantially similar to the second voltage value of the second threshold voltage V−. The first and second threshold voltages V+, V− establish the negative and positive input voltage variations at which the feedforward control signal (or more particular, respective feedforward control signals NT, PT) transitions to indicate a negative or positive input voltage variation, respectively.

The first comparator 280 is coupled to current mirror output node 240 c (e.g., a level-shifted voltage) at a first comparator input (e.g., a non-inverting input) and is also coupled to receive the first threshold voltage V+ at a second comparator input (e.g., an inverting input) and is configured to generate a first comparison signal NT at an output 200 c. The first comparison signal NT, which corresponds to a first feedforward control signal, may be indicative of a negative transient of the input voltage V_(IN) when the current mirror output voltage is less than the first threshold voltage.

The second comparator 290 is coupled to the current mirror output node 240 c at a first comparator input (e.g., an inverting input) and is also coupled to receive the second threshold voltage V− at a second comparator input (e.g., a non-inverting input) and is configured to generate a second comparison signal PT at an output 200 d. The second comparison signal PT, which corresponds to a second feedforward control signal, may be indicative of a positive transient of the input voltage V_(IN) when the current mirror output voltage is greater than the second threshold voltage.

In operation, when a positive input voltage transient occurs, causing a feedforward current I_(FFWD) to flow into current mirror node 240 a and the voltage at node 240 a to rise, the current mirror output voltage V_(RAMP) will fall and when V_(RAMP) hits the threshold voltage V−, the output 200 d of comparator 290 will trip generating logic bit PT to indicate the positive input voltage transient. Conversely, when a negative input voltage transient occurs, causing a feedforward current I_(FWWD) to flow out of current mirror node 240 a and the voltage at node 240 a to fall, the current mirror output voltage V_(RAMP) will rise and when V_(RAMP) hits the threshold voltage V+, the output 200 c of comparator 280 will trip generating logic bit NT to indicate the negative input voltage transient.

Mismatches between the first and second current mirror paths 241, 244 may lead to drift in the current mirror output voltage V_(RAMP) over time. Accordingly, control circuit 200 may include a reset circuit 1210 to counteract this issue. The reset circuit 1210, which is illustrative of one example configuration of a reset circuit, includes a logic gate 1212, a second buffer 1214 and a switch 1216.

The logic gate 1212 is coupled to receive the first comparison signal NT at a first logic gate input 1212 a, the second comparison signal PT at a second logic gate input 1212 c, and a reset signal at a third logic gate input 1212 b and is configured to generate an output signal at logic gate, which signal is used to control operation of the switch 1216, as will be further discussed below.

Switch 1216 is coupled between the current mirror output 240 c and an output of buffer 1214, as shown. The buffer 1214 is coupled to receive a third threshold voltage Vnom and is configured to reset the current mirror output voltage at node 240 c to the threshold voltage Vnom when switch 1216 is closed.

The reset circuit 1210 is configured to periodically reset the current mirror output voltage at node 240 c to a predetermined level, here Vnom. In one example configuration, the switch 1216 is closed to reset the current mirror output voltage after a predetermined number of cycles of a system clock signal (in response to the Reset signal) and also after the occurrence of positive logic bits PT and negative logic bits NT by operation of the OR gate 1212. Example operation of a control circuit according to the disclosure (e.g., 200, shown in FIG. 2) and the reset functionality is described further below in connection with FIG. 3. As will be apparent, because the current mirror output voltage level V_(RAMP) is reset at periodic intervals, certain, relatively slower input voltage variations may not be detected since the voltage on capacitor 250 may not be able to reach the threshold voltage V+ and V− before it is reset to Vnom. However, generally, the feedback path 130 (FIG. 1) is designed to detect such slower input voltage transients.

It is to be appreciated that the feedforward control circuit 200 described above is but one of many potential configurations of feedforward control circuits in accordance with the concepts, systems, circuits and techniques described herein. As one example, while the digitizing circuit 260 of the control circuit 200 is shown and described herein as including first and second comparators 280, 290 to generate respective comparison signals indicative of negative and positive transients of the input voltage V_(IN), respectively, in some embodiments the digitizing circuit 260 can alternatively include more than two comparators and/or other circuitry (e.g., multi-bin ADCs), for example, to detect more subtle changes in the input voltage V_(IN) (and the current mirror output voltage V_(RAMP) to which the digitizing circuit 260 is responsive).

Referring to FIG. 3, illustrative signal waveforms of various input and output signals of a control circuit, which can be the same as or similar to feedforward control circuit 200 shown in FIG. 2, are shown in a plurality of plots (305, 310, 315, 320, 325) having a horizontal axis with a scale in time units and a vertical axis with a scale in units of volts (V). Plot 305 includes a signal 306 representative of input voltage V_(IN) and plot 310 includes a signal 311 representative of a current mirror output voltage V_(RAMP) as may be generated at a current mirror output node (e.g., node 240 c, shown in FIG. 2). Plot 315 includes a signal 316 representative of a reset signal as may be provided to an input 1212 b of logic gate 1212 (FIG. 2) of the control circuit, plot 320 includes a signal 321 representative of a feedforward control signal NT indicative of a negative transient of the input voltage VIN, and plot 325 includes a signal 326 representative of a feedforward control signal PT indicative of a positive transient of the input voltage V_(IN).

As illustrated, signal levels of the current mirror output voltage 311 vary in response to a signal level of the input voltage V_(IN) signal 306. As is also illustrated, the current mirror output voltage 311 is periodically reset in response to the reset signal 316 when there are no input voltage transients present, shown in plot 315.

More particularly, during a first time period to, the input voltage V_(IN) signal 306 is at a first steady state voltage level (i.e., there is no transient). Since no feedforward current I_(FFWD) flows through capacitor 230 under this steady state input voltage condition, current mirror output voltage 311 also remains substantially constant. Additionally, after a predetermined amount of time during the first time period t0 in which no transient in the input voltage V_(IN) signal 306 is detected by the control circuit, a reset pulse 316 a of the reset signal 316 resets the current mirror output voltage 311 to a predetermined level, such as Vnom (FIG. 2).

During a second time period tl, the input voltage V_(IN) signal 306 decreases from the first voltage level to a second voltage level that is substantially less the first voltage level. In response thereto, the current mirror output voltage 311 increases, causing the comparator 280 (FIG. 2) to trip and generate the negative transient signal 321 including a plurality of negative logic bits NT 321 a indicative of the negative transient of the input voltage V_(IN) 306. More particularly, each time the comparator 280 trips, generating an NT pulse or logic bit, the V_(RAMP) voltage 311 is reset (via the NT input to logic gate 1212 of FIG. 2). It will be appreciated that the number of negative logic bits NT provides an indication of the magnitude of the negative input voltage transient.

During a third time period t2, the input voltage V_(IN) signal 306 remains substantially constant, thereby resulting in no feedforward current flow In through capacitor 230 and no change in the voltage across capacitor 250 (i.e., V_(RAMP)). After a predetermined amount of time during the third time period t2 in which no transient in the input voltage V_(IN) signal 306 is detected by the control circuit, reset pulses 316 b of the reset signal 316 reset the current mirror output voltage 311 to a predetermined voltage level, such as Vnom (FIG. 2).

During a fourth time period t3, the input voltage VIN signal 306 increases from the second voltage level to a third voltage level that is substantially greater than the second voltage level. In response thereto, current mirror output voltage 311 decreases, causing the comparator 290 (FIG. 2) to trip and generate the positive transient signal 326 including a plurality of positive logic bits PT 326 a indicative of the positive transient of the input voltage V_(IN) signal 306. More particularly, each time the comparator 290 trips, generating a PT pulse or logic bit, the V_(RAMP) voltage 311 is reset (via the PT input to logic gate 1212 of FIG. 2). It will be appreciated that the number of positive logic bits PT provides an indication of the magnitude of the positive input voltage transient.

During a fifth time period t4, the input voltage V_(IN) signal 306 remains substantially constant, again resulting in no feedforward current flow I_(FFWD) through capacitor 230 and no change in the voltage across capacitor 250 (i.e., V_(RAMP)). Additionally, after a predetermined amount of time during the fifth time period t4 in which no transient in the input voltage V_(IN) signal 306 is detected by the control circuit, reset pulses 316 c of the reset signal 316 reset the current mirror output voltage 311 to a predetermined voltage level, such as V_(nom) (FIG. 2).

According to a further aspect, in embodiments, the PID controller 182 (FIG. 1) may contain an adaptive compensation feature. Referring to FIG. 4, a small signal control loop representation 400 of an embodiment of the Buck regulator 100 including a digital compensator with adaptive compensation is shown. The control loop 400 includes a control to output transfer function 406, Gvd(s), an input voltage to output transfer function 402, Gvg(s), and a term 410 representing the output impedance of the regulator Zout(s). A phase delay inherent to the digital control loop 400 is labelled 422, e^(−st) ^(sw) and the control loop 400 further includes a transfer function of the PID controller 426, Gc(s). The feedback division ratio (ratio associated with the feedback divider circuit 140 of FIG. 1) is assumed to be 1 for simplicity.

The control to output transfer function Gvd(s) given by equation (1):

$\begin{matrix} {{Gvd} = \frac{Vin}{\frac{s^{2}}{\omega_{0}^{2}} + \frac{s}{Q_{eff}\omega_{0}} + 1 + \frac{R_{L}}{R_{OUT}}}} & (1) \end{matrix}$

contains two poles at the corner frequency ω₀ of the LC output filter including inductor L_(x) and a capacitor C_(BP) (FIG. 1). The Q_(eff) in the transfer function is given by (Q_(L)∥Q_(OUT)) where

$Q_{L} = {{\frac{1}{R_{L}}\sqrt{\frac{L_{X}}{C_{OUT}}}\mspace{14mu} {and}\mspace{14mu} Q_{OUT}} = {R_{OUT}{\sqrt{\frac{C_{OUT}}{L_{X}}}.}}}$

Thus, at light loads, the Q_(eff) is dominated by R_(L) which represents conduction losses in the circuit due inductor DC and AC resistance, MOSFET on resistance and other resistive losses. At heavy loads, the Q_(eff) is dominated by low output resistance (R_(OUT)).

The PID transfer function 426 Gc(s) contains two zeros to compensate for the double poles of the control to output transfer function 406, Gvd(s), one pole at DC for infinite gain at DC, and another pole to keep the compensator bounded. The general form of a PID transfer function 426 Gc(s) is shown in equation (2), where ω_(z1) and ω_(z2) are the two zeros and ω_(p) is the pole.

$\begin{matrix} {{Gc} = \frac{{K\left( {\frac{s}{\omega_{z\; 1}} + 1} \right)}\left( \frac{s}{\omega_{z\; 2} + 1} \right)}{s\left( {\frac{s}{\omega_{p}} + 1} \right)}} & (2) \end{matrix}$

To achieve adaptive compensation, several relationships between certain control loop parameters are established and maintained. In particular, the corner frequency ω₀ of the output filter has a first fixed, predetermined relationship with respect to the switching frequency ω_(SW), a second fixed, predetermined relationship with respect to the crossover frequency ω_(C), and a third fixed, predetermined relationship with respect to the at least one pole oop and at least one zero ω_(Z1), ω_(Z2). One example set of such predetermined relationships is give in equation (3):

$\begin{matrix} {{\omega_{c} = {5\omega_{0}}}{\omega_{z\; 1} = \frac{\omega_{0}}{10}}{\omega_{z\; 2} = \omega_{0}}{\omega_{p} = {10\omega_{0}}}{\omega_{SW} = {100\omega_{0}}}} & (3) \end{matrix}$

As will become apparent, the specific predetermined relationships (i.e., ratios) set forth in equation (3) can be varied to suit design requirements; however, once the relationships are fixed, the compensation parameters (i.e., proportional, integral, and derivative gain terms Kp, Ki, Kd, respectively, discussed below) will automatically vary, or adapt as output voltage V_(OUT), switching frequency, and input voltage V_(IN) are varied.

Using the relationships set forth in equation (3) above, the transfer function Gc(s) 426 of the PID controller (i.e., equation (2)) can be simplified as follows:

$\begin{matrix} {{Gc} = \frac{{K\left( {\frac{10s}{\omega_{0}} + 1} \right)}\left( \frac{s}{\omega_{0} + 1} \right)}{s\left( {\frac{s}{10\omega_{0}} + 1} \right)}} & (4) \end{matrix}$

The total loop gain of the converter is given by Gvd×Gc. At the crossover frequency ω_(c), the total loop gain is set to 1 and phase margin is given by: π+

(Gvd (j ω_(c))×Gc (j ω_(c)). Since the crossover frequency ω_(c) is set at five times the output filter corner frequency wo, the phase of the converter (i.e.,

(Gvd (s=j ω_(c))) can be estimated to be approximately −180° (presuming that the Q_(eff) is not too high). More particularly, the phase of the converter Gvd (i.e.,

(Gvd (s=j ω_(c))) can be approximately given by

$\tan^{- 1}{\frac{5}{24Q_{eff}}.}$

since tnis converter phase is very small for relatively small Q_(eff), the phase margin of the system is simply the phase gain provided by compensator at the crossover frequency

(Gc (s=j ω_(c)); which by solving equation (4) is approximately 76°. Thus, for Q_(eff)>0.72, the phase of the compensator is approximately 76°. In case of Q_(EFF)<0.72, the phase margin would be higher than 76° because the phase lag due to double poles of LC filter would be less.

For the magnitude computation, it assumed that due to double poles at the LC corner frequency ω₀ the control to output transfer function Gvd will drop by 40 dB/decade from the corner frequency. Hence, the magnitude of the transfer function Gvd (s=j ω_(c)) ˜V_(IN)/25. It will be appreciated that this expression is based on the relationships of equation (3) and specifically based on having the crossover frequency at approximately five times the LC corner frequency.

Based on this same relationship of the crossover frequency being approximately five times the corner frequency and equation (5), it follows that the magnitude of the PID controller transfer function Gc (s=j ω_(c))=45.62×K/ω₀.

Because it is desirable to keep the total loop gain (i.e., Gc×Gvd) approximately equal to one at the crossover frequency, it is desirable to maintain the following expression, from which K can be determined:

$\begin{matrix} {{45.62\frac{K}{\omega_{0}}\frac{V_{IN}}{25}} = 1} & (5) \end{matrix}$

From equation (5), it can be seen that the compensator DC gain K is inversely proportional to the input voltage V_(IN). This relationship forms the basis of the described adaptive compensation techniques. As illustrated below, for digital compensation, the proportional, integral and derivative parameters only depend on input voltage V_(IN) and their relationship to switching frequency is eliminated due to the relationships set forth in equation (3).

Referring back to equation (4), in order to convert this analog compensator expression into the discrete domain, bilinear transformation

$\left( s\rightarrow{\frac{2}{t_{sw}}\left( \frac{z - 1}{z + 1} \right)} \right)$

can be used which preserves the phase and gain matching between the analog and digital compensators up to half the switching frequency. Upon applying bilinear transformation, the discrete counterpart of Gc can be expressed by equation (6):

$\begin{matrix} {{{Gcd} = {\frac{{Kt}_{sw}}{2\pi}\frac{\left( {{10^{3}\left( {z - 1} \right)} + {\pi \left( {z + 1} \right)}} \right)\left( {{10^{2}\left( {z - 1} \right)} + {2{\pi \left( {z + 1} \right)}}} \right)}{\left( {z - 1} \right)\left( {{10\left( {z - 1} \right)} + {\pi \left( {z + 1} \right)}} \right)}}}{{Gcd} = {\frac{{Kt}_{sw}}{2\pi}\left( {A_{0} + \frac{A_{1}z}{z - 1} + {A_{2}\frac{z - 1}{z - \alpha}}} \right)}}} & (6) \end{matrix}$

Substituting the value of K from equation (5), the discrete counterpart of Gc can be expressed by equation (7):

$\begin{matrix} {{Gcd} = {\frac{1}{180.48V_{IN}}\left( {A_{0} + \frac{A_{1}z}{z - 1} + {A_{2}\frac{z - 1}{z - \alpha}}} \right)}} & (7) \end{matrix}$

Comparing equation (7) with the PID equation of a discrete compensator, it will be appreciated that the proportional compensator gain term Kp is given by A₀/180.48 V_(IN), the integral compensator gain term Ki is given by A₁/180.48 V_(IN), and the derivative compensator gain term Kd is given by A₂/180.48 V_(IN). Consideration of each of these compensator gain parameters reveals that the compensator gain is not dependent on switching frequency and is inversely proportional to input voltage V_(IN). As a result, the same compensator parameters Kp, Ki, Kd can be used while the switching frequency is varied as long as the ratio between the corner frequency (fixed by proper selection of LC filter) and switching frequency is maintained.

Referring also to FIG. 5, an example PID controller 182 implementing adaptive compensation is shown to receive the feedback control signal e[n] from the first compensator input 180 a (FIG. 1) at a PID controller input 182 a and is configured to generate a first duty cycle word d_(PID)[n] associated with the feedback control signal e[n] at a PID controller output 182 b. More particularly, the controller 182 includes a proportional portion 500, an integral portion 510 and a derivative portion 520. As will be explained, each of the controller portions 500, 510, 520 includes a respective gain term Kp, Ki, Kd stored in a respective register 504, 518, and 526, which gain terms are adaptively adjusted by an adaptive compensation unit 550 described below.

The proportional portion 500 includes a multiplier 502 configured to multiply a stored proportional gain Kp 504 by the feedback control signal e[n] to generate a proportional term 500a (Kp x e[n]) that is proportional to the error e[n]. The proportional term 500 a is coupled to a summation element 540, as shown.

Integral portion 510 of the controller 182 includes a multiplier 516, the feedforward circuit 184 (FIG. 1) in embodiments in which feedforward control is implemented, an integrating or integral register 514, and a delay element 512. Multiplier 516 is configured to multiply the stored integral gain Ki 518 by the feedback control signal e[n]. Integrating register 514 is responsive to the duty cycle word d_(FFwd)[n] associated with the received first or second feedforward signals. Thus, in some embodiments, the compensator module 186 (FIG. 1) may be part of the PID controller 182 in which case the duty cycle word d_(FFwd)[n] may be coupled directly into the PID controller 182. The integral term 510 a at the output of the integrating register 514 can be given by Ki(Σe[n])+d_(FFwd)[n] and is proportional to the integral of the error e[n]. The integral term 510 a is coupled to the summation element 540. It will be appreciated that in embodiments not implementing feed forward control, the integral term is given simply by Ki(Σe[n]).

The derivative portion 520 of the controller 182 includes a multiplier 524 configured to multiply the stored derivative gain Kd 526 by a difference between a current duty cycle error e[n] and a previous duty cycle error e[n−1] as generated by elements 528, 530. The output of the multiplier 524 can thus be expressed as Kd(e[n]−e[n−1]). Difference element 530 generates a derivative term 520 a that is proportional to the derivative of the error e[n] by subtracting the term Kd(e[n]−e[n−1]) from a previous value of the derivative term 520 a multiplied by a gain a 536 as implemented with elements 534, 538

Summation element 540 is responsive to the proportional term 500 a, the integral term 510 a, and the derivative term 520 a to generate the duty cycle word d_(PID)[n] associated with the feedback control signal e[n]. As noted above, PWM circuit 190 (FIG. 1) is configured to generate a PWM, or voltage control signal having a duty cycle based on the duty cycle word d[n].

The adaptive compensation unit 550 is responsive to the integral term 510 a at the output of the integrating register 514 and generates and adaptively adjusts compensator gain parameters Kp 504, Ki 518, Kd 528. Advantageously, the integral term 510 a holds information about the input voltage V_(IN) because, even when the error term e[n] goes to zero, the integral term 510 a keeps integrating the error value and so, holds the duty cycle associated with the input voltage. A relatively large duty cycle word in the integrating register 514 corresponds to a relatively small input voltage V_(IN) and conversely, a smaller duty cycle word in the integrating register 514 corresponds a larger input voltage V_(IN). This is especially true for a system where output voltage V_(OUT) is fixed but other parameters like the switching frequency or input voltage V_(IN) are varied.

In some applications, the output voltage V_(OUT) (FIG. 1) may be changed by using a different resistance ratio for the feedback divider circuit 140 (FIG. 1). Significantly, in such embodiments in which the resistor divider ratio is varied to vary the output voltage V_(OUT), the compensator gain parameters Kp 504, Ki 518, Kd 528 scale accordingly because the total loop gain includes the feedback divider circuit. Thus, even though a larger output voltage V_(OUT) would generally require larger compensator gain parameters Kp 504, Ki 518, Kd 528 to achieve the same phase margin for the same crossover frequency, because the larger output voltage results in a larger duty cycle, this larger duty cycle is captured in the integrating register 514, which register value in turn proportionately scales the compensator gain parameters Kp 504, Ki 518, Kd 528. Furthermore, use of the integrating register value to scale the compensator gain parameters Kp 504, Ki 518, Kd 528 also captures variations in duty cycle attributable to input voltage V_(IN) variations. Stated differently, a larger duty cycle would produce proportionally larger compensator gain parameters. A larger duty cycle is also indicative of smaller input and consequently larger compensator gain parameters. The adaptive compensation unit 550 that operates to scale the compensator gain parameters Kp 504, Ki 518, Kd 528 in a manner directly proportional to the stored duty cycle word thus adjusts the gain parameters in a manner that accounts for variations in feedback divider ratio implemented to vary the output voltage V_(OUT) and also for variations in the input voltage V_(IN). In embodiments, the adaptive compensation unit 550 can re-compute the compensator gain parameters Kp 504, Ki 518, Kd 528 in digital cycles when duty cycle computation is not being performed, thus adding little computational overhead to entire system.

Referring to FIG. 6, example waveforms associated with operation of the PID controller 182 (FIG. 5) are shown. In particular, an example input voltage Vin is labelled 604. Feedforward signals are labelled 608, 610 and can represent the first feedforward signal PT 608 indicative of a positive transient of the input voltage V_(IN) and a second feedforward signal NT 610 indicative of a negative transient of the input voltage V_(IN). The resulting regulated output voltage V_(OUT) is labelled 616.

The variation of output voltage 616 in response to the input voltage transients is better (i.e., less variation) than if fixed gains were used for the gain parameters Kp 504, Ki 518, and Kd 526. In other words, adaptive compensation unit 550 provides some improvement in transient response performance. However according to FIG. 7, additional methodologies can be used to achieve further transient performance benefits.

Notably, the digital compensator 180 (FIG. 1) computes the duty cycle only once per regulator switching cycle and the switching frequency of the regulator tends to be significantly slower than the frequency of the digital compensator (180 of FIG. 1). As a result, if the PT and NT signals occur multiple times during a switching cycle, the digital compensator would not be able to keep up with the input voltage transients. As a result, the even with the transient response improvement of waveform 604 (as compared to fixed gain terms without adaptive control), there is a noticeable lag in the output voltage correction.

The circuitry and techniques described below in connection with FIG. 7 result in a further transient response improvement as is shown by output voltage waveform 620 which has smaller variation than waveform 616. Suffice it to say here that this additional performance improvement can be achieved by generating a feedforward gain value Kffw (which is used to scale a feedforward control signal) in response to the square of a value indicative of the input voltage as will be described. Further performance benefits can be achieved by updating the feedforward gain value each time the value indicative of the input voltage is updated. As an alternative, the feedforward control signals can be accumulated and the feedforward gain value can be updated less frequently (e.g., once per switching clock cycle) based on the accumulated feedforward control signals.

Referring to FIG. 7, an alternative digital compensator, or controller 700 is shown to receive the feedback control signal e[n] from the first compensator input 180 a (FIG. 1) at a controller input 700 a and is configured to generate a compensator duty cycle word d[n] at a controller output 700 c. The controller 700 incorporates PID control and includes a proportional portion 702, an integral portion 710 and a derivative portion 740. Each of the controller portions 702, 710, 740 includes a respective gain term Kp, Ki, Kd stored in a respective register 706, 718, and 746, which gain terms are adaptively adjusted by an adaptive compensation unit 730. The controller 700 further incorporates feedforward control and includes a feedforward circuit 724 that is coupled to receive feedforward control signals UP, DN at a controller inputs 700 b and an adaptive feedforward compensation unit 720 for the feedforward control.

The proportional portion 702 includes a multiplier 704 configured to multiply a stored proportional gain Kp 706 by the feedback control signal e[n] to generate a proportional term 702 a (Kp x e[n]) that is proportional to the error e[n]. The proportional term 702 a is coupled to a summation element 770, as shown.

Integral portion 710 of the controller 700 includes a multiplier 716, feedforward circuit 724, an integrating or integral register 714, and a delay element 712. Multiplier 716 is configured to multiply the stored integral gain Ki 718 by the feedback control signal e[n]. The integral register 710 receives the output of multiplier 716 and also a feedforward control signal scaled by a feedforward gain value Kffw, as shown. The integral term 710 a (labeled Ireg) at the output of the integrating register 714 is proportional to the integral of the error e[n] and thus is indicative of the input voltage V_(IN) and is proportional to the duty cycle. The integral term 710 a is coupled to the summation element 770. As will be explained, the adaptive feedforward compensation unit 720 operates to generate the feedforward gain value Kffw.

The derivative portion 740 of the controller 700 includes a multiplier 744 configured to multiply the stored derivative gain Kd 746 by a difference between a current duty cycle error e[n] and a previous duty cycle error e[n−1] as generated by elements 748, 750. The output of the multiplier 744 can thus be expressed as Kd(e[n]−e[n−1]). Difference element 752 generates a derivative term 740 a that is proportional to the derivative of the error e[n] by subtracting the term Kd(e[n]−e[n−1]) from a previous value of the derivative term multiplied by a gain a 756 as implemented with elements 754, 758.

Summation element 770 is responsive to the proportional term 702 a, the integral term 710 a, and the derivative term 740 a to generate the duty cycle word d[n] associated with the feedback control signal e[n]. As noted above, PWM circuit 190 (FIG. 1) is configured to generate a PWM, or voltage control signal having a duty cycle based on the duty cycle word d[n].

The adaptive compensation unit 730 is responsive to the integral term 710 a at the output of the integrating register 714 and generates and adaptively adjusts compensator gain parameters Kp 706, Ki 718, Kd 746. Advantageously, the integral term 710 a holds information about the input voltage V_(IN) because, even when the error term e[n] goes to zero, the integral term 710 a keeps integrating the error value and so, holds the duty cycle associated with the input voltage. A relatively large duty cycle word in the integrating register 714 corresponds to a relatively small input voltage V_(IN) and conversely, a smaller duty cycle word in the integrating register 714 corresponds a larger input voltage V_(IN). This is especially true for a system where output voltage V_(OUT) is fixed but other parameters like the switching frequency or input voltage V_(IN) are varied.

In some applications, the output voltage V_(OUT) (FIG. 1) may be changed by using a different resistance ratio for the feedback divider circuit 140 (FIG. 1). Significantly, in such embodiments in which the resistor divider ratio is varied to vary the output voltage V_(OUT), the compensator gain parameters Kp 706, Ki 718, Kd 746 scale accordingly because the total loop gain includes the feedback divider circuit. Thus, even though a larger output voltage V_(OUT) would generally require larger compensator gain parameters Kp 706, Ki 718, Kd 746 to achieve the same phase margin for the same crossover frequency, because the larger output voltage results in a larger duty cycle, this larger duty cycle is captured in the integrating register 714, which register value in turn proportionately scales the compensator gain parameters Kp 706, Ki 718, Kd 746.

Use of the integrating register value 714 to scale the compensator gain parameters Kp 706, Ki 718, Kd 746 captures variations in duty cycle attributable to input voltage V_(IN) variations. Stated differently, a larger duty cycle would produce proportionally larger compensator gain parameters. A larger duty cycle is also indicative of smaller input and consequently larger compensator gain parameters. The adaptive compensation unit 730 that operates to scale the compensator gain parameters Kp 706, Ki 718, Kd 746 in a manner directly proportional to the stored duty cycle word thus adjusts the gain parameters in a manner that accounts for variations in feedback divider ratio implemented to vary the output voltage V_(OUT) and also for variations in the input voltage V_(IN). In embodiments, the adaptive compensation unit 730 can re-compute the compensator gain parameters Kp 706, Ki 718, Kd 746 in digital cycles when duty cycle computation is not being performed, thus adding little computational overhead to entire system. The adaptive compensation unit 730 can operate at the regulator switching frequency fsw.

The adaptive feedforward compensation unit 720 is coupled to receive the Ireg value 710 a that is indicative of the input voltage and generates the feedforward gain value Kffw in response to the square of the Ireg value (Ireg²). The feedforward circuit 724 scales the feedforward control signals UP, DN by the feedforward gain value Kffw to generate a feedforward input to the integral register 714, as shown.

The following equation can be used to define the necessary duty cycle change (Kffw) in response to a change in the input voltage ΔV_(IN) (i.e., when either of the feedforward inputs PT, NT) is detected.

$\begin{matrix} {{\partial D} = {\frac{\partial V_{OUT}}{\partial V_{IN}} = {{\frac{- D^{2}}{V_{OUT}}\Delta \; V_{IN}} = {- K_{ffw}}}}} & (8) \end{matrix}$

where ∂D can be the compensator output duty cycle word d[n] in FIG. 1. Every time a PT or NT pulse is detected, it indicates that the input voltage has moved by a known AVmi value and therefore the duty cycle must be moved by the quantity Kffw which represents the feedforward gain in the circuit. The duty cycle value stored in the integral register 714 allows feedforward gain to scale as duty cycle is varied. Higher input voltage leads to lower duty cycle for a fixed output voltage. At higher input voltage, lower feedforward gain is desired to avoid overcompensation. Stated differently, for a known ΔV_(IN), the duty cycle should change by Kffw. For higher input voltage, if the input voltage moves by a fixed amount, then the amount of adjustment needed to the duty cycle is small; whereas, for lower input voltage, if the input voltage moves by that same fixed amount, the amount of adjustment needed to the duty cycle is larger.

As noted above, the switching frequency of the regulator fsw tends to be significantly slower than frequency fcontroller at which the digital compensator, or controller 700 operates. As a result, the feedforward control signals UP, DN can occur more often than the duty cycle is updated by summation element 770. Here, the benefits of the adaptive feedforward control achieved by the adaptive feedforward compensation unit 720 can be enhanced by updating the feedforward gain value Kffw more often than the duty cycle d[n] is updated. For example, the feedforward gain value Kffw can be updated every time the Ireg value 710 a is updated. In some instances, the Ireg value 710 a is updated any time one of the positive feedforward control signal UP or the negative feedforward control signal DN is received and also once per each switching cycle (e.g., at the end of a clock cycle corresponding to the switching frequency when the duty cycle word d[n] is updated). By updating the Ireg value 710 a any time a feedforward control signal UP, DN is received and also once every switching cycle, the compensator signal (i.e., duty cycle word d[n]) will cause the regulated output voltage V_(OUT) to respond more effectively to input voltage transients than otherwise possible.

Updating the feedforward gain value Kffw more often than just once every switching cycle is particularly advantageous because of the squared relationship between the Ireg value 710 a and the feedforward gain Kffw as illustrated by equation (8) above. In other words, the squared relationship results in the feedforward gain value Kffw being even more affected by the Ireg value than if it were just a proportional relationship.

Referring again to FIG. 6, waveform 620 illustrates the regulated output voltage V_(OUT) generated by a regulator incorporating the controller 700 of FIG. 7 in response to the input voltage waveform 604 and associated feedforward control signals 608, 610. As is apparent, the transient response performance of output voltage 620 is better than the performance illustrated by waveform 616 which waveform 616 corresponds to use of the controller 182 of FIG. 5 that does not incorporate the adaptive feedforward compensation unit 720.

Alternatively, the feedforward circuit 724 can operate to accumulate the positive feedforward control signals UP and negative feedforward control signals DN occurring during a switching clock cycle and the accumulated signals UP, DN can be applied all at once to update the compensator duty cycle word d[n] once per switching cycle. By allowing the controller 700 to accumulate all the UP and DN signals received in a switching cycle, the controller provides improved response to faster line transients.

Referring also to FIG. 8, an alternative feedforward control circuit 800 for generating a feedforward control signal is shown. The control circuit 800, which may provide the feedforward path 170 in the regulator circuit 100 of FIG. 1, may be used in place of the control circuit 200 (FIG. 2) in some instances. The control circuit 800 receives input voltage V_(IN) at an input 800 a and generates a feedforward control signal based on the input voltage V_(IN). In the circuit 800, the feedforward control signal may take the form of a first control signal UP provided at a first feedforward output 800 b and/or a second control signal DN provided at a second feedforward output 800 c. Feedforward control signals UP, DN can provide the inputs 700 b to the controller in FIG. 7. As will be apparent, signals UP, DN can be similar to signals PT, NT of FIG. 2; however, signals UP, DN are indicative of the direction (i.e., polarity) of a change in the input voltage VIN (i.e., up or down) whereas signals PT, NT are indicative of the occurrence of a predetermined transient in the input voltage (i.e., positive transient or negative transient). It will be appreciated that the feedforward control circuits 200 (FIG. 2), 800 (FIG. 8) are example control circuits that can be used in the controller of FIG. 5 and/or the controller of FIG. 7.

The input voltage V_(IN) is coupled to a voltage to current converter 804 that generates a current 806 proportional to the input voltage. A current DAC 810 is configured to generate a current 812 proportional to a count value of a counter 816 that represents the value of the input voltage. Currents 806 and 812 (both of which can be in binary digital form) can be compared by elements 820 and 822 to determine whether the input voltage V_(IN) is increasing or decreasing. Elements 820, 822 can include hysteresis so that the outputs only switch when the input moves up or down by a predetermined input voltage amount. If the input voltage V_(IN) is increasing, then the counter 816 is incremented; whereas if the input voltage V_(IN) is decreasing, then the counter 816 is decremented. Counter 816 generates the UP feedforward control signal at output 800 b as may take the form of a pulse when the input voltage V_(IN) is increasing and further generates the DN feedforward control signal at output 800 c as may take the form of a pulse occurring when the input voltage V_(IN) is decreasing.

As described above and as will be appreciated by those of ordinary skill in the art, embodiments of the disclosure herein may be configured as a system, method, or combination thereof. Accordingly, embodiments of the present disclosure may be comprised of various means including hardware, software, firmware or any combination thereof.

Having described preferred embodiments, which serve to illustrate various concepts, structures and techniques, which are the subject of this patent, it will now become apparent to those of ordinary skill in the art that other embodiments incorporating these concepts, structures and techniques may be used. Additionally, elements of different embodiments described herein may be combined to form other embodiments not specifically set forth above. For example, while the regulator 100 of FIG. 1 is described as including both the feedforward path and associated circuitry described further in connection with FIGS. 2 and 3 and also as including the adaptive compensation controller 182 described further in connection with FIGS. 4 and 5, it will be appreciated that regulators may benefit from incorporation of either of these features individually as well.

Accordingly, it is submitted that that scope of the patent should not be limited to the described embodiments but rather should be limited only by the spirit and scope of the following claims. 

What is claimed is:
 1. A control circuit for a voltage regulator comprising at least one switch responsive to a voltage control signal for switching at a switching frequency, the voltage regulator configured to convert an input voltage into a regulated output voltage, comprising: a divider coupled to the regulated output voltage to generate a divided voltage having a value that is a fraction of the regulated output voltage; an ADC responsive to the divided voltage to generate a feedback control signal; a digital compensator responsive to the feedback control signal and to a feedforward control signal scaled by a feedforward gain value to generate a compensator signal, wherein the digital compensator comprises a register configured to store a value indicative of the input voltage and a feedforward gain unit configured to generate the feedforward gain value in response to the value indicative of the input voltage; and a pulse width modulator responsive to the compensator signal to generate the voltage control signal.
 2. The control circuit of claim 1, wherein the digital compensator further comprises a proportional-integral-derivative (PID) controller, wherein the register is an integral register and wherein one or more of a proportional gain, an integral gain, and a derivative gain of the PID controller is scaled by the value indicative of the input voltage.
 3. The control circuit of claim 1, wherein the feedforward gain unit is configured to generate the feedforward gain value in response to the square of the value indicative of the input voltage.
 4. The control circuit of claim 1, wherein the feedforward control signal comprises one or more of positive feedforward control signals and negative feedforward control signals.
 5. The control circuit of claim 4, wherein the feedforward gain value is updated in response to the value indicative of the input voltage being updated.
 6. The control circuit of claim 5, wherein the feedforward gain value is updated each time the value indicative of the input voltage is updated.
 7. The control circuit of claim 6, wherein the value indicative of the input voltage is updated (1) each time one of the positive feedforward control signals and negative feedforward control signals is received; and (2) at an end of a clock cycle corresponding to the switch frequency.
 8. The control circuit of claim 4, wherein the digital compensator further comprises an accumulator configured to accumulate the positive feedforward control signals and negative feedforward control signals occurring during a clock cycle corresponding to the switching frequency, wherein the compensator signal is updated at the end of the clock cycle based on the positive feedforward control signals and negative feedforward control signals accumulated during the clock cycle.
 9. The control circuit of claim 2, wherein the digital compensator is configured to generate the compensator signal during a first portion of a clock cycle corresponding to the switching frequency and the gain of one or more of a proportional gain, an integral gain, and a derivative gain of the PID controller is scaled by the value indicative of the input voltage during a second portion of the clock cycle different than the first portion of the clock cycle.
 10. The control circuit of claim 1, wherein the voltage regulator is a DC-DC converter.
 11. A method for generating a voltage control signal for controlling a switch of a voltage regulator configured to convert an input voltage into a regulated output voltage, the switch operating at a switching frequency, comprising: generating a feedback control signal based on a voltage difference between the regulated output voltage and a reference voltage; generating a compensator signal with a digital compensator in response to the feedback control signal and in response to a feedforward control signal scaled by a feedforward gain value, wherein the feedforward control signal comprises one or more of positive feedforward control signals and negative feedforward control signals; and converting the compensator signal into the voltage control signal with a pulse width modulator.
 12. The method of claim 11, wherein generating the compensator signal comprises providing a digital compensator with a Proportional-Integral-Derivative (PID) controller and wherein the method further comprises scaling one or more of a proportional gain, an integral gain, or a derivative gain of the PID controller by the value indicative of the input voltage.
 13. The method of claim 12, wherein providing the digital compensator with a PID controller comprises providing an integral register in which a value indicative of the input voltage is stored and wherein the method further comprises computing the feedforward gain value in response to the value indicative of the input voltage.
 14. The method of claim 13, wherein computing the feedforward gain value comprises computing the feedforward gain value in response to the square of the value indicative of the input voltage.
 15. The method of claim 13, wherein computing the feedforward gain value comprises updating the feedforward gain value each time the value indicative of the input voltage is updated.
 16. The method of claim 15, wherein the value indicative of the input voltage is updated (1) each time one of the positive feedforward control signal or negative feedforward control signal is received; and (2) at an end of a clock cycle corresponding to the switch frequency.
 17. The method of claim 11, wherein the method further comprises accumulating the positive feedforward control signals and negative feedforward control signals occurring during a clock cycle corresponding to the switching frequency, wherein the compensator signal is updated at the end of the clock cycle based on the positive feedforward control signals and negative feedforward control signals accumulated during the clock cycle.
 18. A control circuit for a voltage regulator comprising at least one switch responsive to a voltage control signal for switching at a switching frequency, the voltage regulator configured to convert an input voltage into a regulated output voltage, comprising: a divider coupled to the regulated output voltage to generate a divided voltage having a value that is a fraction of the regulated output voltage; an ADC responsive to the divided voltage to generate a feedback control signal; means, responsive to the feedback control signal and to a feedforward control signal scaled by a feedforward gain value, for generating a compensator signal; and a pulse width modulator responsive to the compensator signal to generate the voltage control signal.
 19. The control circuit of claim 18, wherein the compensator signal generating means comprises means, responsive to a value indicative of the input voltage, for generating the feedforward gain value.
 20. The control circuit of claim 19, wherein the feedforward gain value generating means comprises means for updating the feedforward gain value when the value indicative of the input voltage is updated. 